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rev. 0 information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a AD8628 one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781/329-4700www.analog.com fax: 781/326-8703 ? analog devices, inc., 2002 zero-drift, single-supply rail-to-rail input/output operational amplifier pin configurations 5-lead sot-23 (rt suffix) 1 2 3 5 4 ?n +in v+ out AD8628 v 8-lead soic (r suffix) 1 2 3 4 8 7 6 5 AD8628 ?n v +in v+ out nc nc nc nc = no connect features lowest auto-zero amplifier noise low offset voltage: 1 v input offset drift: 0.02 v/ c rail-to-rail input and output swing 5 v single-supply operation high gain, cmrr, and psrr: 120 db very low input bias current: 100 pa low supply current: 1.0 ma overload recovery time: 10 s no external components required applications automotive sensors pressure and position sensors strain gage amplifiers medical instrumentation thermocouple amplifiers precision current sensing photodiode amplifier general description this new breed of amplifier has ultralow offset, drift, and bias current. the AD8628 is a wide bandwidth auto-zero amplifier featuring rail-to-rail input and output swings and low noise. operation is fully specified from 2.7 v to 5 v single supply ( 1.35 v to 2.5 v dual supply). the AD8628 family provides benefits previously found only in expensive auto-zeroing or chopper-stabilized amplifiers. using analog devices?new topology, these zero-drift amplifiers combine low cost with high accuracy and low noise. (no external capacitors are required.) in addition, the AD8628 greatly reduces the digital switching noise found in most chopper-stabilized amplifiers. with an offset voltage of only 1 v, drift less than 0.005 v/ c, and noise of only 0.5 v p-p (0 hz to 10 hz), the AD8628 is perfectly suited for applications where error sources cannot be tolerated. position and pressure sensors, medical equipment, and strain gage amplifiers benefit greatly from nearly zero drift over their operating temperature range. many systems can take advan tage of the rail- to - rail input and output swings provided by the AD8628 family to reduce input biasing complexity and maximize snr. the AD8628 family is specified for the extended industrial tem perature r ange (?0 c to +125 c). the AD8628 amplifier is ava il abl e in the tiny sot-23 and the popular 8-lead narrow soic plastic packages. the sot-23 package devices are avail able only in tape and reel.
?2? rev. 0 AD8628especifications electrical characteristics parameter symbol conditions min typ max unit input characteristics offset voltage v os 15 v e40 c t a +125 c10 v input bias current i b 30 100 pa e40 c t a +125 c 1.5 na input offset current i os 50 200 pa e40 c t a +125 c 250 pa input voltage range 05v common-mode rejection ratio cmrr v cm = 0 v to 5 v 120 140 db e40 c t a +125 c 115 130 db large signal voltage gain * a vo r l = 10 k , v o = 0.3 v to 4.7 v 125 145 db e40 c t a +125 c 120 135 db offset voltage drift v os / t e40 c t a +125 c 0.002 0.02 v/ c output characteristics output voltage high v oh r l = 100 k to ground 4.99 4.996 v e40 c t a +125 c 4.99 4.995 v r l = 10 k to ground 4.95 4.98 v e40 c t a +125 c 4.95 4.97 v output voltage low v ol r l = 100 k to v+ 1 5 mv e40 c t a +125 c25mv r l = 10 k to v+ 10 20 mv e40 c t a +125 c1520mv short circuit limit i sc 25 50 ma e40 c t a +125 c 40 ma output current i o 30 ma e40 c t a +125 c 15 ma power supply power supply rejection ratio psrr v s = 2.7 v to 5.5 v e40 c t a +125 c 115 130 db supply current/amplifier i sy v o = 0 v 0.85 1.1 ma e40 c t a +125 c 1.0 1.2 ma input capacitance differential c in 1.5 pf common mode 10 pf dynamic performance slew rate sr r l = 10 k 1.0 v/ s overload recovery time 0.05 ms gain bandwidth product gbp 2.5 mhz noise performance voltage noise e n p-p 0.1 hz to 10 hz 0.5 v p-p e n p-p 0.1 hz to 1.0 hz 0.16 v p-p voltage noise density e n f = 1 khz 22 nv/ hz hz hz
?3? rev. 0 AD8628 electrical characteristics parameter symbol conditions min typ max unit input characteristics offset voltage v os 15 v e40 c t a + 125 c10 v input bias current i b 30 100 pa e40 c t a + 125 c 1.0 1.5 na input offset current i os 50 200 pa e40 c t a + 125 c 250 pa input voltage range 05v common-mode rejection ratio cmrr v cm = 0 v to 2.9 v 115 130 db e40 c t a + 125 c 110 120 db large signal voltage gain a vo r l = 10 k , v o = 0.3 v to 4.7 v 110 140 db e40 c t a + 125 c 105 130 db offset voltage drift v os / t e40 c t a + 125 c 0.002 0.02 v/ c output characteristics output voltage high v oh r l = 100 k to ground 2.68 2.695 v e40 c t a + 125 c2 .68 2.695 v r l = 10 k to ground 2.67 2.68 v e40 c t a + 125 c2 .67 2.675 v output voltage low v ol r l = 100 k to v+ 1 5 mv e40 c t a + 125 c25mv r l = 10 k to v+ 10 20 mv e40 c t a + 125 c1520mv short circuit limit i sc 10 15 ma e40 c t a + 125 c 10 ma output current i o 10 ma e40 c t a + 125 c 5ma power supply power supply rejection ratio psrr v s = 2.7 v to 5.5 v e40 c t a + 125 c 115 130 db supply current/amplifier i sy v o = 0 v 0.75 1.0 ma e40 c t a + 125 c 0.9 1.1 ma input capacitance differential c in 1.5 pf common mode 10 pf dynamic performance slew rate sr r l = 10 k 1v/ s overload recovery time 0.05 ms gain bandwidth product gbp 2 mhz noise performance voltage noise e n p-p 0.1 hz to 10 hz 0.5 v p-p voltage noise density e n f = 1 khz 22 nv/ hz hz hz
AD8628 ?4? rev. 0 absolute maximum ratings 1 supply voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 v input voltage . . . . . . . . . . . . . . . . gnd e 0.3 v to v se + 0.3 v differential input voltage 2 . . . . . . . . . . . . . . . . . . . . . . 5.0 v output short-circuit duration to gnd . . . . . . . . . indefinite storage temperature range r, rt packages . . . . . . . . . . . . . . . . . . . . e65 c to +150 c operating temperature range . . . . . . . . . . e40 c to +125 c junction temperature range r, rt packages . . . . . . . . . . . . . . . . . . . . e65 c to +150 c lead temperature range (soldering, 60 sec) . . . . . . . . 300 c 1 stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sec- tions of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 differential input voltage is limited to 5 v or the supply voltage, whichever is less. package type ja * jc unit 5-lead sot-23 (rt) 230 146 c/w 8-lead soic (r) 158 43 c/w * ja is specified for worst-case conditions, i.e., ja is specified for device soldered in circuit board for surface-mount packages. caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the AD8628 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. warning! esd sensitive device ordering guide temperature package package branding model range description option information AD8628art e40 c to +125 c 5-lead sot-23 rt-5 aya AD8628ar e40 c to +125 c 8-lead soic r-8
AD8628 ?5? rev. 0 input offset voltage e v e2.5 2.5 e1.5 e0.5 0.5 1.5 number of amplifiers 180 160 0 80 60 40 20 140 100 120 v s = 2.7v t a = 25 c tpc 1. input offset voltage distribution at 2.7 v input offset voltage e v e2.5 2.5 e1.5 e0.5 0.5 1.5 number of amplifiers 100 80 0 40 30 20 10 70 50 60 v s = 5v v cm = 2.5v t a = 25 c 90 tpc 4. input offset voltage distribution at 5 v load current e ma 1 0.01 0.001 output voltage e mv 0.1 1 10 0.1 10 1k 100 0.0001 0.01 v s = 2.7v source sink tpc 7. output voltage to supply rail vs. load current at 2.7 v input common-mode voltage e v input bias current e pa 60 0 01 6 23 45 +85 c 40 30 20 10 +25 c e40 c 50 v s = 5v tpc 2. input bias current vs. input common-mode voltage at 5 v tcvos e nv/ c number of amplifiers 7 0 01 0 2 468 6 5 4 3 2 1 v s = 5v t a = e40 c to +125 c tpc 5. input offset voltage drift temperature e c input bias current e pa 1,500 1,150 0 e50 e25 175 02550 100 125 150 75 800 450 100 v s = 5v v cm = 2.5v t a = e40 c to +150 c tpc 8. input bias current vs. temperature input common-mode voltage e v input bias current e pa 1,500 e1,500 01 6 23 45 150 c 500 0 e500 e1,000 1,000 125 c v s = 5v tpc 3. input bias current vs. input common-mode voltage at 5 v load current e ma 1 0.01 0.001 output voltage e mv 0.1 1 10 0.1 10 1k 100 0.0001 0.01 v s = 5v t a = 25 c source sink tpc 6. output voltage to supply rail vs. load current at 5 v temperature e c supply current e a 1,250 1,000 0 e50 200 050 100 150 750 500 250 t a = 25 c 5v 2.7v tpc 9. supply current vs. temperature t ypical performance characteristicse
AD8628 ?6? rev. 0 supply voltage e v supply current e a 1,000 0 800 200 600 400 01 6 2345 t a = 25 c tpc 10. supply current vs. supply voltage frequency e hz closed-loop gain e db 50 e30 1k 10k 100k 1m 10m 30 20 10 0 e10 40 e20 70 60 a v = 100 a v = 10 a v = 1 v s = 2.7v c l = 20pf r l = 2k tpc 13. closed-loop gain vs. frequency at 2.7 v frequency e hz output impedance e 100 1k 100m 10k 100k 10m 300 270 0 240 210 180 150 120 90 60 30 v s = 5v a v = 100 a v = 1 a v = 10 1m tpc 16. output impedance vs. frequency at 5 v frequency e hz open-loop gain e db 10k 100k 1m 10m 70 60 30 50 40 30 20 10 0 10 20 0 45 90 135 180 225 phase shift e de g rees v s = 2.7v c l = 20pf r l = m = 52.1 tpc 11. open-loop gain and phase vs. frequency frequency e hz closed-loop gain e db 1k 10m 10k 100k 1m a v = 100 a v = 10 a v = 1 v s = 5v c l = 20pf r l = 2k 50 e30 30 20 10 0 e10 40 e20 70 60 tpc 14. closed-loop gain vs. frequency at 5 v time e 4 s/div voltage e 500mv/div 0 0 0 00 0 00000000 0 0 0 0 0 0 v s = 1.35v c l = 300pf r l = a v = 1 tpc 17. large signal transient response at 2.7 v frequency e hz open-loop gain e db 10k 100k 1m 10m 70 60 30 50 40 30 20 10 0 10 20 45 90 135 180 225 0 phase shift e de g rees v s = 5v c l = 20pf r l = m = 52.1 tpc 12. open-loop gain and phase vs. frequency frequency e hz output impedance e 100 1k 100m 10k 100k 10m 300 270 0 240 210 180 150 120 90 60 30 v s = 2.7v a v = 100 a v = 1 a v = 10 1m tpc 15. output impedance vs. frequency at 2.7 v time e 5 s/div voltage e 1v/div 0 0 0 00 0 00000000 0 0 0 0 0 0 v s = 2.5v c l = 300pf r l = a v = 1 tpc 18. large signal transient response at 5 v
AD8628 ?7? rev. 0 time e 4 s/div voltage e 50mv/div 0 0 0 00 0 00000000 0 0 0 0 0 0 v s = 1.35v c l = 50pf r l = a v = 1 tpc 19. small signal transient response at 2.7 v capacitive load e p f o vershoot e % 110 1k 100 80 0 70 60 50 40 30 20 10 os+ ose v s = 2.5v r l = 2k t a = 25 c tpc 22. small signal overshoot vs. load capacitance at 5 v time e 200 s/div voltage e 1v/div 0 0 0 00 0 00000000 0 0 0 0 0 0 v s = 2.5v v in = 1khz @ 3v p-p c l = 0pf r l = 10k a v = 1 tpc 25. no phase reversal time e 4 s/div voltage e 50mv/div 0 0 0 00 0 00000000 0 0 0 0 0 0 v s = 2.5v c l = 50pf r l = a v = 1 tpc 20. small signal transient response at 5 v time e 2 s/div voltage e v 0 0 0 00 0 00000000 0 0 0 v out 0 0 v s = 2.5v a v = e50 r l = 10k c l = 0 ch1 = 50mv/div ch2 = 1v/div 0v v in 0v tpc 23. positive overvoltage recovery frequency e hz cmrr e db 100 1k 10 m 10k 100k 1m 140 120 e60 100 80 60 40 20 0 e20 e40 v s = 2.7v tpc 26. cmrr vs. frequency at 2.7 v capacitive load e pf o vershoot e % 110 1k 100 100 90 0 80 70 60 50 40 30 20 10 os+ ose v s = 1.35v r l = 2k t a = 25 c tpc 21. small signal overshoot vs. load capacitance at 2.7 v time e 10 s/div voltage e v 0 0 0 00 0 00000000 0 0 0 0 0 0 v s = 2.5v a v = e50 r l = 10k c l = 0 ch1 = 50mv/div ch2 = 1v/div 0v v in 0v v out tpc 24. negative overvoltage recovery frequency e hz 100 1k 10m 10k 100k 1m v s = 5v cmrr e db 140 120 e60 100 80 60 40 20 0 e20 e40 tpc 27. cmrr vs. frequency at 5 v
AD8628 ?8? rev. 0 frequency e hz 100 1k 10m 10k 100k 1m v s = 1.35v + psrr psrr psrr e db 140 120 e60 100 80 60 40 20 0 e20 e40 tpc 28. psrr vs. frequency frequency e hz output swing e v p-p 5.5 4.5 0 100 1k 1m 10k 100k 3.5 2.5 0.5 1.5 5.0 4.0 3.0 2.0 1.0 v s = 5v r l = 10k t a = 25 c a v = 1 tpc 31. maximum output swing vs. frequency at 5 v voltage noise density e nv/ hz chz s sahz c d hzhz chz s s s s c s la s c hzhz lasds hz chz s sahz c d hzhz chz s s l a c a c s la s c hzhz lasds hz c hz s sahz c d hzhz
AD8628 ?9? rev. 0 frequency e khz 120 75 0 025 5101520 105 90 45 15 60 30 voltage noise density e nv/ hz s sahz c d hzhz a c shccca s a c c sc sc c scc a c alla l cc h l cc h l cc h s c chz s lasds hz c a c shccca s a c c sc sc c scc a c slc s a c c c s a c alla l cc h l cc h l cc h s c
AD8628 ?10? rev. 0 functional description the AD8628 is a single-supply, ultrahigh precision rail-to-rail input and output operational amplifier. the typical offset voltage of less than 1 v allows this amplifier to be easily configured for high gains without risk of excessive output voltage errors. the extremely small temperature drift of 5 nv/ c ensures a minimum of offset voltage error over its entire temperature range of e40 c to +125 c, making the AD8628 amplifier ideal for a variety of sensitive measurement applications in harsh operating environments. the AD8628 achieves a high degree of precision through a patented combination of auto-zeroing and chopping. this unique topology allows the AD8628 to maintain its low offset voltage over a wide temperature range and over its operating lifetime. AD8628 also optimizes the noise and bandwidth over previous generations of auto-zero amplifiers, offering the lowest voltage noise of any auto-zero amplifier by more than 50%. previous designs used either auto-zeroing or chopping to add precision to the specifications of an amplifier. auto-zeroing results in low noise energy at the auto-zeroing frequency at the expense of higher low frequency noise due to aliasing of wideband noise into the auto-zeroed frequency band. chopping results in lower low frequency noise at the expense of larger noise energy at the chopping frequency. AD8628 uses both auto-zeroing and chopping in a patented ping-pong arrangement to obtain lower low frequency noise together with lower energy at the chopping and auto-zeroing frequencies, maximizing the signal-to-noise ratio (snr) for the majority of applications without the need for additional filtering. the relatively high clock frequency of 15 khz simplifies filter requirements for a wide, useful, noise-free bandwidth. AD8628 is one of the few auto-zero amplifiers offered in the 5-lead sot-23 package. it greatly improves the ac parameters of the previous auto-zero amplifiers. it has low noise over a relatively wide bandwidth (0 hz to 10 khz) and can be used where the highest dc precision is required. in systems with signal bandwidths up to 5 khz to 10 khz, the AD8628 provides true 16-bit accuracy making it the best choice for very high resolution systems. 1/f noise 1/f noise, also known as pink noise, is a major contributor of errors in dc-coupled measurements. this 1/f noise error term can be in the range of several v or more, and, when amplified with the closed-loop gain of the circuit, can show up as a large output offset. for example, when an amplifier with a 5 v p-p 1/f noise is configured for a gain of 1,000, its output will have 5 mv of error due to the 1/f noise. but AD8628 eliminates 1/f noise internally and therefore greatly reduces output errors. here is how it works: 1/f noise appears as a slowly varying offset to AD8628 inputs. auto-zeroing corrects any dc or low frequency offset, thus the 1/f noise component is essentially removed, leaving AD8628 free of 1/f noise. frequency e khz 120 105 0 012 4 vo ltag e noise density e nv/ hz ad hz) lc hz) l c hz) ahzallahs sdadc ad z hz hz hzhz z
AD8628 ?11? rev. 0 AD8628 peak-to-peak noise vs. competition because of the ping-pong action between auto-zeroing and chopping, the peak-to-peak noise of the AD8628 is much lower than its competition. figures 2 and 3 show this comparison. time e 1s/div 0 0 0 00 0 vo ltag e e 0.5 v/div 00000000 0 0 0 0 0 0 e n p-p = 0.5 v bw = 0.1hz to 10hz figure 2. AD8628 peak-to-peak noise time e 1s/div 0 0 0 00 0 vo ltag e e 0.5 v/div 00000000 0 0 0 0 0 0 e n p-p = 2.3 v bw = 0.1hz to 10hz figure 3. ltc2050 peak-to-peak noise noise behavior with first order low-pass filter AD8628 was simulated as a low-pass filter and then configured as shown in figure 4. the behavior of the AD8628 matches the simulated data. it was verified that noise is rolled off by first order filtering. in out 470pf 100k 1k figure 4. test circuit: first order low-pass filter: x101 gain and 3 khz corner frequency frequency e hz 100 10 20 30 40 50 60 70 80 90 45 20 noise e db 30 10 5 25 15 40 35 50 figure 5a. simulation transfer function of test circuit frequency e khz 45 20 noise e db 30 10 5 25 15 40 35 50 figure 5b. actual transfer function of test circuit measured noise spectrum of test circuit showing noise between 5 khz and 45 khz is successfully rolled off by first order filter. total integrated input-referred noise for first order filter (AD8628 vs. competition) 3db filter bandwidth e hz 10 1 0.1 10 10k 100 rms noise e v 1k lt c2050 ad8551 AD8628 figure 6. 3 db filter bandwidth in hz for a first order filter, the total integrated noise from the AD8628 is lower than the ltc2050.
AD8628 ?12? rev. 0 input overvoltage protection although the AD8628 is a rail-to-rail input amplifier, care should be taken to ensure that the potential difference between the inputs does not exceed the supply voltage. under normal negative feedback operating conditions, the amplifier will correct its output to ensure the two inputs are at the same voltage. however, if either input exceeds either supply rail by more than 0.3 v, large currents will begin to flow through the esd protection diodes in the amplifier. these diodes are connected between the inputs and each supply rail to protect the input transistors against an electrostatic discharge e vent and are normally reverse biased. however, if the input voltage exceeds the supply voltage, these esd diodes will become forward biased. without current limiting, excessive amounts of current could flow through these diodes, causing permanent damage to the device. if inputs are subject to overvoltage, appropriate series resistors should be inserted to limit the diode current to less than 5 ma maximum. output phase reversal output phase reversal occurs in some amplifiers when the input common-mode voltage range is exceeded. as common-mode voltage is moved outside of the common-mode range, the outputs of these amplifiers will suddenly jump in the opposite direction to the supply rail. this is the result of the differential input pair shutting down, causing a radical shifting of internal voltages that results in the erratic output behavior. the AD8628 amplifier has been carefully designed to prevent any output phase reversal, provided both inputs are maintained within the supply voltages. if one or both inputs could exceed either supply voltage, a resistor should be placed in series with the input to limit the current to less than 5 ma. this will ensure the output will not reverse its phase. overload recovery time many auto-zero amplifiers are plagued by long overload recovery time, often in milliseconds, due to the complicated settling behavior of the internal nulling loops after saturation of the outputs. AD8628 has been designed so that internal settling occurs w ithin two clock cycles after output saturation happens. this results in a much shorter recovery time, less than 10 s, when compared to other auto-zero amplifiers. the wide bandwidth of the AD8628 enhances performance when it is used to drive loads that inject transients into the outputs. t his is a common situation when an amplifier is used to drive the input of switched capacitor adcs. time e 500 s/div 0 0 0 00 0 vo ltag e e v 00000000 0 0 0 0 0 0 v in 0v 0v v out ch 1 = 50mv/div ch 2 = 1v/div a v = e50 figure 7. positive input overload recovery for AD8628 time e 500 s/div 0 0 0 00 0 vo ltag e e v 00000000 0 0 0 0 0 0 v in 0v 0v v out ch 1 = 50mv/div ch 2 = 1v/div a v = e50 figure 8. positive input overload recovery for ltc2050 time e 500 s/div 0 0 0 00 0 vo ltag e e v 00000000 0 0 0 0 0 0 v in 0v 0v v out ch 1 = 50mv/div ch 2 = 1v/div a v = e50 figure 9. positive input overload recovery for lmc2001
AD8628 ?13? rev. 0 time e 500 s/div 0 0 0 00 0 vo ltag e e v 00000000 0 0 0 0 0 0 v in 0v 0v v out ch 1 = 50mv/div ch 2 = 1v/div a v = e50 figure 10. negative input overload recovery for AD8628 time e 500 s/div 0 0 0 00 0 vo ltag e e v 00000000 0 0 0 0 0 0 v in 0v 0v v out ch 1 = 50mv/div ch 2 = 1v/div a v = e50 figure 11. negative input overload recovery for ltc2050 time e 500 s/div 0 0 0 00 0 vo ltag e e v 00000000 0 0 0 0 0 0 v in 0v 0v v out ch 1 = 50mv/div ch 2 = 1v/div a v = e50 figure 12. negative input overload recovery for lmc2001 the results shown in figures 7-12 are summarized in table i. table i. overload recovery time product type positive overload negative overload recovery recovery ( s) recovery ( s) AD8628 6 9 ltc2050 650 25,000 lmc2001 40,000 35,000 infrared sensors infrared (ir) sensors, particularly thermopiles, are increasingly being used in temperature measurement for applications as wide-ranging as automotive climate controls, human ear thermometers, home- insulation analysis, and automotive repair diagnostics. the relatively small output signal of the sensor demands high gain with very low offset voltage and drift to avoid dc errors. if interstage ac coupling is used (figure 13), low offset and drift prevents the input amplifier?s output from drifting close to saturation. the low input bias currents generate minimal errors from the sensor?s output impedance. as with pressure sensors, the very low amplifier drift with time and temperature eliminates additional errors once the temperature measurement has been calibrated. the low 1/f noise improves snr for dc measurements taken over periods often exceeding 1/5 second. figure 15 shows a circuit that can amplify ac signals from 100 v to 300 v microvolts up to the 1 v to 3 v level, gain of 10,000 for accurate a/d conversion. 5v 100k 10k 5v 100 v e 300 v 100 to bias vo ltag e 10k f c 1.6hz ir detector 100k 10 f AD8628 AD8628 figure 13. preamplifier for thermopile
AD8628 ?14? rev. 0 precision current shunts a precision shunt current sensor benefits from the unique attributes of auto-zero amplifiers when used in a differencing configuration (figure 14). shunt current sensors are used in precision current sources for feedback control systems. they are also used in a variety of other applications, including battery fuel gauging, laser diode power measurement and control, torque feedback controls in electric power steering, and precision power metering. r s 0.1 supply i r l 100 100k c 5v 100 100k c e = 1,000 r s i 100mv/ma AD8628 figure 14. low-side current sensing in such applications, it is desirable to use a shunt with very low resistance to minimize the series voltage drop; this minimizes wasted power and allows the measurement of high currents with- out saving power. a typical shunt might be 0.1 . at measured current values of 1 a, the shunt?s output signal is hundreds of millivolts, or even volts, and amplifier error sources are not criti- cal. however, at low measured current values in the 1 ma range, the 100 v output voltage of the shunt demands a very low offset voltage and drift to maintain absolute accuracy. low input bias currents are also needed, so that injected bias current does not become a significant percentage of the measured current. high open-loop gain, cmrr, and psrr all help to maintain the overall circuit accuracy. as long as the rate of change of the current is not too fast, an auto-zero amplifier can be used with excellent results. output amplifier for high precision dacs AD8628 is used as an output amplifier for a 16-bit high precision dac in unipolar configuration. in this case, the selected op amp needs to have very low offset voltage (the dac lsb is 38 v when operated with a 2.5 v reference) to eliminate the need for outp ut offset trims. input bias current (typically a few tens of pico amp) must also be very low since it generates an additional zero code error when multiplied by the dac output impedance (approxi- mately 6 k ). rail-to-rail input and output provides full-scale output with very little error. output impedance of the dac is constant and code-independent, but the high input impedance of the AD8628 minimizes gain errors. the amplifier?s wide bandwidth also serves well in this case. the amplifier with settling time of 1 s adds another time constant to the system, increasing the settling time of the output. t he settling time of the ad5541 is 1 s. the combined settling time is approximately 1.4 s, as can be derived from the equation: t total t dac t ad sss () = () + () 22 8628 10 f ref(ref * ) v dd refs * cs d scl ldac dd ad la ad adl sal ac adad ada
AD8628 C15C rev. 0 outline dimensions dimensions shown in millimeters 5-lead plastic surface-mount package [sot-23] (rt-5) 2.90 pin 1 1.60 bsc 2.80 bsc 1.95 bsc 0.95 bsc 1 3 4 5 2 0.22 0.08 0.60 0.45 0.30 10 0 0.50 0.30 0.15 max seating plane 1.45 max coplanarity 1.30 1.15 0.90 compliant to jedec standards mo-178aa dimensions shown in millimeters and (inches) 8-lead standard small outline package (r-8) 0.25 (0.0098) 0.19 (0.0075) 1.27 (0.0500) 0.40 (0.0157) 0.50 (0.0196) 0.25 (0.0099) 45 8 0 1.75 (0.0688) 1.35 (0.0532) seating plane 0.25 (0.0098) 0.10 (0.0040) 85 4 1 5.00 (0.1968) 4.80 (0.1890) pin 1 4.00 (0.1574) 3.80 (0.1497) 1.27 (0.0500) bsc 6.20 (0.2440) 5.80 (0.2284) 0.51 (0.0201) 0.33 (0.0130) coplanarity controlling dimensions are in millimeters; inch dimensions (in parentheses) are rounded-off millimeter equivalents for reference only and are not appropriate for use in design compliant to jedec standards ms-012 aa
C16C rev. 0 c02735C0C5/02(0) printed in u.s.a. C16C

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